INTEGRATED CIRCUITS
AN1777
Low voltage front-end circuits:
SA601, SA620
M. B. Judson
1997 Aug 20
Philips
Semiconductors
Philips Semiconductors
Application note
Low voltage front-end circuits: SA601, SA620
Author: M. B. Judson
CONTENTS
I.
Introduction . . . . . . . . . . . . . . . . . . . . . . . . .
II.
Key Attributes of the SA601 and SA620
III. Power Consumption . . . . . . . . . . . . . . . . . .
IV. Low Noise Amplifiers . . . . . . . . . . . . . . . . .
V.
SA620 Mixer . . . . . . . . . . . . . . . . . . . . . . . . .
Open-Collector Output Basics . . . . . . . . . . . . . . . . . . . . .
Why R
C
Acts Like A Source Resistor . . . . . . . . . . . . . . .
Open-collector with R
LOAD
. . . . . . . . . . . . . . . . . . . . . . . .
Open-collector with inductor (L
C
) . . . . . . . . . . . . . . . . . .
Open-collector with Inductor (L
C
) and R
LOAD
. . . . . . . .
AN1777
2
2
2
3
3
4-46
4-46
4-46
4-48
4-51
implementing a high loaded-Q
µ-strip
inductor and an extensive
discussion of open-collector mixer outputs and how to match them
will also be presented.
The SA601 and SA620 are products designed for high performance
low power RF communication applications from 800 to 1200MHz.
These chips offer the system designer an alternative to discrete
front-end designs which characteristically introduce a great deal of
end-product variation, require external biasing components, and
require a substantial amount of LO drive. The SA601 and SA620
contain a low noise amplifier and mixer, offering an increase in
manufacturability and a minimum of external biasing components
due to integration. The LO drive requirements for active mixers are
less stringent than passive mixers, thus minimizing LO isolation
problems associated with high LO drive-levels. These chips also
contain power down circuitry for turning off all or portions of the chip
while not in use. This minimizes the average power consumed by
the front-end circuitry. The SA620 features an internal VCO that
eliminates additional cost and space needed for an external VCO.
The SA601 and SA620 fit within a 20-pin surface mount plastic
shrink small outline package (SSOP), thus saving a considerable
amount of space.
VI. Flexible Matching Circuit . . . . . . . . . . . . .
VII. SA601 MIXER . . . . . . . . . . . . . . . . . . . . . . . .
VIII. Matching the Open-Collector Differential
Output . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Inductor L
2
. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Capacitors C
5
and C
7
. . . . . . . . . . . . . . . . . . . . . . . . .
Inductor L
3
. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Capacitor C
6
. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Resistor R
2
. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Method of Achieving High Impedance Matching with a
Network Analyzer . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Non-Ideal Current-Combiner Ckt Considerations . . . . .
9
10
11
4-54
4-54
4-54
4-54
4-54
4-54
4-55
II.
KEY ATTRIBUTES OF THE SA601 AND SA620
IX.
Matching Examples . . . . . . . . . . . . . . . . . .
83MHz, 1 kΩ Match . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
45MHz, 1 kΩ Match . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
110.592MHz, 50Ω Match . . . . . . . . . . . . . . . . . . . . . . . . .
SA601 IP3
IN
Considerations . . . . . . . . . . . . . . . . . . . . . .
Summary of Mixer Open-Collector Output Concepts . .
14
4-56
4-58
4-59
4-61
4-62
The primary differences between the SA601 and SA620 are the LNA
power down capability, the implementation of the mixer output
circuitry, and the incorporation of an integrated VCO.
Table 1 below summarizes the attributes of both parts.
X.
SA601 Mixer Characterization . . . . . . . . .
SA601 System SINAD Performance . . . . . . . . . . . . . . .
20
4-62
XI. SA620 VCO . . . . . . . . . . . . . . . . . . . . . . . . . .
XII.
µ-Strip
Inductor Oscillator
Resonant Circuit . . . . . . . . . . . . . . . . . . . . .
General Theoretical Background . . . . . . . . . . . . . . . . . . .
Increasing Loaded-Q . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
High-Q Short Microstrip Inductor . . . . . . . . . . . . . . . . . . .
UHF VCO Using the SA620 at 900MHz . . . . . . . . . . . . .
20
21
4-63
4-64
4-64
4-65
XIII. Application Boards . . . . . . . . . . . . . . . . . . .
26
SA601 Applications Board . . . . . . . . . . . . . . . . . . . . . . . . 4-68
SA601 Application Board Modification For Increasing Mixer
Gain . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4-68
SA620 Applications Board . . . . . . . . . . . . . . . . . . . . . . . . 4-69
Noise Figure and Gain . . . . . . . . . . . . . . . . . . . . . . . . . . .
1dB Compression Point . . . . . . . . . . . . . . . . . . . . . . . . . .
Input Third-Order Intercept Point . . . . . . . . . . . . . . . . . . .
Phase Noise . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
4-71
4-72
4-73
4-73
XV. Common Questions and Answers . . . . .
XVI. References . . . . . . . . . . . . . . . . . . . . . . . . . .
I.
INTRODUCTION
31
31
The objectives of this application note are to highlight key features
and distinguish key differences between the SA601 and SA620.
The power, gain, noise figure, and third-order intercept point of the
LNA and mixer will be characterized. A resonant circuit
1995 Aug 10
2
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XIV. Test and Measurement Tips . . . . . . . . . . .
29
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Product
SA601
SA620
Diff. Mixer
Output
Yes
No
LNA Thru
Mode
No
Mixer
Power
Down
Yes
Yes
Int. VCO
and VCO
Pwr Down
No
Yes
Yes
Table 1. Showing SA601 and SA602 Attributes
III.
POWER CONSUMPTION
As mentioned above, the average power consumed by the front-end
circuitry can be decreased by selectively turning off circuitry that is
not in use. The supply current at a given voltage will decrease more
than 3mA for each LNA, mixer, or VCO disabled. When the LNA is
disabled on the SA620 it is replaced by a 9dB attenuator. This is
useful for extending the dynamic range of the receiver when an
overload condition exists. Tables 2 and 3 below contain averaged
data taken on the SA601 and SA620 while in an application board
environment.
Table 2. Showing SA601 Supply Current
V
CC
3.0
4.0
5.0
I
CC
(mA)
8.4
8.4
8.3
I
CC
(mA) Mixer Disabled
4.9
4.7
4.5
Table 3. Showing SA620 Supply Current
V
CC
3.0
4.0
5.0
I
CC
(mA)
11.4
11.6
11.7
I
CC
(mA)
LNA
Disabled
8.0
8.5
8.9
I
CC
(mA)
Mixer
Disabled
8.1
8.0
8.0
I
CC
(mA)
VCO
Disabled
8.2
8.2
8.2
I
CC
(mA)
Chip Fully
Powered
Down
1.4
1.6
1.7
Philips Semiconductors
Application note
Low voltage front-end circuits: SA601, SA620
AN1777
14
12
MAGNITUDE (dB)
10
8
6
4
2
0
860
868
876
884
892
900
908
916
924
932
940
Gain (dB)
VCC = 3, 4 and 5V
IV.
LOW NOISE AMPLIFIERS
Noise Figure (dB)
VCC = 3, 4 and 5V
The performance of the SA601 and SA620 low noise amplifiers are
virtually identical. You can expect an average gain of approximately
11.5
±1.5dB
and a noise figure of approximately 1.6
±0.3dB.
The
LNA input and output networks are matched for optimum return loss,
gain and best noise figure over the 869 - 894MHz band. They also
perform well when utilized in the 902 - 928MHz band without any
additional modification to the LNA matching networks. Figures 1
and 2 show gain and noise figure of a typical SA601 and SA620
LNA in the application board environment. Both the gain and the
noise figure remain almost constant as V
CC
is adjusted to 3, 4 and
5V. Therefore, only one curve is shown for clarity.
SR01275
FREQUENCY (MHz)
Figure 1. Noise Figure and Gain vs Frequency SA601 LNA
14
12
10
MAGNITUDE (dB)
8
6
4
2
0
860
868
876
884
892
900
908
916
924
932
940
Gain (dB)
VCC = 3, 4 and 5V
V.
SA620 MIXER
Noise Figure (dB)
VCC = 3, 4 and 5V
FREQUENCY (MHz)
SR01276
Figure 2. Noise Figure and Gain vs Frequency SA620 LNA
The SA620 mixer is intended for operation with the integrated VCO
and employs a single open-collector output structure. The
open-collector output structure allows the designer to easily match
any high impedance load for maximum power transfer with a
minimum of external components. This eliminates the need for
elaborate matching networks. The external mixer output circuitry
also incorporates a network which distributes the power from the
mixer output to two unequal loads. This enables the mixer output to
be matched to a high impedance load such as a SAW bandpass
filter (typically 1kΩ) while simultaneously providing a 50Ω test point
that can be used for production diagnostics. The mixer output
circuitry generates the majority of questions for those utilizing this
part in their current applications, so some basic concepts regarding
open-collector outputs are presented below, as well as a discussion
of the network used to provide the 50Ω diagnostic point.
V
CC
R
C
R
BB
v
i
V
BB
+
–
v
i
+
–
R
BB
B
+
v
be
–
E
C
g
m
v
be
R
C
C
g
m
v
be
R
C
V
T
+
–
R
C
r
π
Basic Transistor
AC Equivalent Model
AC Equivalent Output
Thevenin’s Equivalent
Circuit
Symbol Convention:
DC: represented by uppercase symbols with uppercase subscripts, i.e., I
C
AC: represented by lowercase symbols with lowercase subscripts, i.e., i
C
Combined AC & DC: represented by lowercase symbol with uppercase subscript, i.e., i
C
SR00001
Figure 3. R
C
as Source Resistor
1995 Aug 10
3
Philips Semiconductors
Application note
Low voltage front-end circuits: SA601, SA620
AN1777
DC Model
AC Model
V
CC
i
C
R
C
•
•
•
Kill AC sources
Open capacitors
Short inductors
V
CC
R
C
I
C
+
v
CE
–
v
i
+
–
•
•
•
Kill DC sources
Short capacitors
Open inductors
R
C
R
BB
+
v
CE
–
R
BB
+
v
CE
+
V
BE
–
–
V
BB
R
BB
v
i
+
–
i
B
V
BB
I
+
*
1
R
C
Eq. 1.
V
)
V
CC
R
C
i
+
*
1
(v )
CE
R
C
C
CE
C
Eq. 2.
SR0002
Figure 4. Basic Transistor Analysis
Open-Collector Output Basics [1, 2, 3, 6]
Why R
C
Acts Like A Source Resistor
An open-collector output allows a designer the flexibility to choose
the value of the R
C
resistor. Choosing this resistor value not only
sets the DC bias point of the device but also defines the source
impedance value. Figure 3 shows the AC model of the transistor.
Converting the output structure by applying a Norton and Thevenin
transformation, one can conclude that R
C
becomes the source
resistance. Thus, by choosing R
C
to be equal to the load, maximum
power transfer will then occur.
Figure 4 shows an active transistor with a collector and base
resistor. From basic transistor theory Equation 1 is generated and
has the same form as the general equation for a straight line
y
+
mx
)
b
The slope of the DC load line is generated by the value of the
collector resistor (m = -1/Rc) and is shown in Figure 5. For a given
small signal base current, the collector current is shown by the
dotted curve.
load (R
LOAD
) and neglect any reactance. Since a resistive load is
used (see Figure 7), the AC output swing is measured at V
OUT
or
V
CE
.
A DC blocking capacitor is used between the R
LOAD
and the V
CE
output to assure that the Q-point is not influenced by R
LOAD
. It is
also necessary to avoid passing DC to the load in applications
where the load is a SAW filter. However, R
LOAD
will affect the AC
load line which is seen in Equation 4 in Figure 7. Notice that the
V
CE
voltage swing is reduced and thus, the V
OUT
signal is reduced
(see Figure 8).
Since the value of R
C
and R
LOAD
affects the AC load line slope, the
value chosen is important. The higher the impedance of R
LOAD
and
Rc, the greater the AC output swing will be at the output, which
means more conversion gain in a mixer. This is due to the slope
getting flatter, thus allowing for more output swing.
i
C
*
1
R
C
V
DC LOAD LINE: SLOPE =
The intersection of the dotted curve and the DC load line is called
the Quiescent point (Q-point) or DC bias determinant. The location
of the Q-point is important because it determines where the
transistor is operated; in the cutoff, active, or saturation regions. In
most cases, the Q-point should be in the active region because this
is where the transistor acts like an amplifier.
Figure 6 shows the ac collector-emitter voltage (V
CE
) output swing
with respect to an AC collector current (i
c
). Collector current is
determined by the AC voltage presented to the input transistor’s
base (v
i
) because it effects the base current (i
b
) which then effects
i
c
. This is how the v
i
is amplified and seen at the output. Recall that
this is with no external load (R
LOAD
) present at the collector. Since
no load is present, the AC load line has an identical slope as the DC
load line as seen in equation 1 and 2 (m = -1/Rc).
CC
R
C
Q-point
I
C
I
B
V
CE
V
CC
v
CE
Open-Collector With R
LOAD
A filter with some known input impedance is a typical load for the
output of the transistor. For simplicity, we will assume a resistive
1995 Aug 10
4
Figure 5. Load Line and Q-Point Graph
SR00003
Philips Semiconductors
Application note
Low voltage front-end circuits: SA601, SA620
AN1777
iC
V
DC LOAD LINE: SLOPE =
CC
R
C
*
1
R
C
iB2
i
C2
Q-point
IB
ic
I
C
i
B1
i
C1
vce
Vce
VCC
vCE
SR00004
Figure 6. Graphical Analysis for the Circuitry in Figure 4
DC Model
AC Model
V
CC
i
C
R
C
•
•
•
Kill AC sources
Open capacitors
Short inductors
V
CC
R
C
I
C
+
v
CE
–
v
i
+
–
•
•
•
Kill DC sources
Short capacitors
Open inductors
R
C
R
BB
+
v
CE
–
+
v
OUT
–
R
LOAD
R
BB
i
B
+
v
CE
+
V
BE
–
–
R
BB
+
v
OUT
–
R
LOAD
V
BB
v
i
+
–
V
BB
I
+
*
1
R
C
Eq. 3.
V
)
V
CC
R
C
i
+
*
1
*
1
)
R
R
C
LOAD
Eq. 4.
(v
)
C
CE
C
CE
SR00005
Figure 7. Basic Transistor Analysis with R
LOAD
1995 Aug 10
5
电源技术
